Prior art current resonant switching power source devices are widely known as having their high power conversion efficiency with low noise and reduced switching loss because a resonance circuit therein converts electric current flowing through switching elements into a sinusoidal wave form for zero current switching (ZCS) when the switching elements are turned on or off. For example, a prior art resonant switching power source device shown in FIG. 6 comprises first and second main MOS-FETs 1 and 2 as first and second switching elements connected in series to a DC power source 3, a series circuit which includes a current resonance capacitor 4, a leakage inductance 5d and a primary winding 5a of a transformer 5 connected in parallel to second main MOS-FET 2, a voltage pseudo resonance capacitor 7 connected between drain and source terminals of second main MOS-FET 2, a first output rectifying diode 8 which has an anode terminal connected to one end of first secondary winding 5b of transformer 5, a first output smoothing capacitor 9 connected between a cathode terminal of first output smoothing capacitor 8 and the other end of first secondary winding 5b, a second output rectifying diode 18 which has an anode terminal connected to one end of second secondary winding 5c of transformer 5, a second output smoothing capacitor 19 connected between cathode terminal of second output rectifying diode 18 and the other end of second secondary winding 5c, and a step down chopper 27 connected to second output smoothing capacitor 19. First output rectifying diode 8 and first output smoothing capacitor 9 are incorporated together to form a first rectifying smoother 10 which produces a first DC output voltage VO1 through first DC output terminals 11 and 12. Second output rectifying diode 18 and second output smoothing capacitor 19 are incorporated together to form a second rectifying smoother 20 which produces second DC output voltage VO2 through step down chopper 27 from second DC output terminals 21 and 22.
Transformer 5 comprises a leakage inductance 5d acting as a current resonance reactor connected equivalently in series to primary winding 6a, and an excitation inductance 5e connected equivalently in parallel to primary winding 6a. Connected to both ends of first output smoothing capacitor 9 in first rectifying smoother 10 is a first output voltage detector 13 for detecting first DC output voltage VO1 issued from first rectifying smoother 10 to produce an error signal VE1, the differential between detected first DC output voltage VO1 and a reference voltage for prescribing the detected output voltage VO1, and error signal VE1 is transmitted to a feedback signal input terminal FB of main control circuit 15 through a photo-coupler 14 of a light emitter 14a and a light receiver 14b. 
Step down chopper 27 comprises a chopping MOS-FET 23 which has a drain terminal connected to a junction of second output rectifying diode 18 and second output smoothing capacitor 19 in second rectifying smoother 20, a flywheel diode 24 connected between a source terminal of chopping MOS-FET 23 and ground terminal 22 in secondary side, a filter reactor 25 which has one end connected to a junction between source terminal of chopping MOS-FET 23 and cathode terminal of flywheel diode 24, and a filter capacitor 26 connected between the other end of filter reactor 25 and ground terminal 22 in secondary side. A chopper controller 28 has a built-in power supply for producing reference voltage to prescribe a value of second output voltage, and produces a PWM signal VS2 of modulated pulse width based on an error signal, the differential between output voltage VO2 on filter capacitor 26 and the reference voltage. Thus, step down chopper 27 controls the on-off operation of chopping MOS-FET 23 by PWM signal VS2 from chopper controller 28 to produce, from second DC output terminals 21 and 22, second DC output VO2 of a constant level lower than DC voltage input into step down chopper 27 from second output smoothing capacitor 19 in second rectifying smoother 20.
As shown in FIG. 7, main control circuit 15 comprises an oscillator 29 for producing pulse signals VPL of frequency variable in response to voltage level of error signal VE1 transmitted from first output voltage detector 13 into feedback signal input terminal FB through photo-coupler 14 of light emitter 14a and light receiver 14b, an inverter 30 for producing an inverted signal of a pulse signal VPL from oscillator 29, a first dead time adder 31 for incorporating a constant dead time in inverted signal VPL from oscillator 29 through inverter 30 to produce a first drive signal VG1, a level shift circuit 32 for shifting voltage level of first drive signal VG1 inclusive of added dead time, a high side buffer amplifier 33 for applying first drive signal VG1 from level shift circuit 32 to a gate terminal of first main MOS-FET 1, a second dead time adder 34 for incorporating a constant dead time in a pulse signal VPL from oscillator 29 to produce a second drive signal VG2, and a low side buffer amplifier 35 for applying second drive signal VG2 inclusive of dead time to a gate terminal of second main MOS-FET 2. Pulse signal VPL has the varied frequency and a constant pulse width, and first drive signal VG1 has a fixed off-period and an on-period varied in response to voltage level of error signal VE1 from output voltage detector 13. Second drive signal VG2 has a fixed on-period and an off-period varied in response to voltage level of error signal VE1 from output voltage detector 13, and therefore, main control circuit 15 produces first and second drive signals VG1 and VG2 to gate terminal of respectively first and second main MOS-FETs 1 and 2. In this way, first and second main MOS-FETs 1 and 2 are alternately turned on and off in response to voltage level of error signal VE1 from first output voltage detector 13.
Referring now to FIG. 8 showing time charts of voltages and electric currents at selected locations in FIG. 6, and the operation of the resonant switching power source device of FIG. 6 is described hereinafter. When second main MOS-FET 2 is turned from on to off at a point t1 in time under the off-condition of first main MOS-FET 1, electric current ICi flows from leakage and excitation inductances 5d and 5e of transformer 5 through voltage pseudo resonance capacitor 7, current resonance capacitor 4 to excitation inductance 5e of transformer 5 while releasing energy accumulated in leakage and excitation inductances 5d and 5e of transformer 5. This causes electric charge in voltage pseudo resonance capacitor 7 to drop voltage VQ1 between drain and source terminals of first main MOS-FET 1 and adversely raise voltage VQ2 between drain and source terminals of second main MOS-FET 2.
When charged voltage in voltage pseudo resonance capacitor 7 comes up to voltage E at a point t2 in time under the off-condition of both first and second main MOS-FETs 1 and 2, voltage VQ1 between drain and source terminals of first main MOS-FET 1 becomes substantially zero and at the same time voltage VQ2 between drain and source terminals of second main MOS-FET 2 becomes substantially equal to voltage E of DC power source 3. During the period of time between points t2 and t3, electric current ICi flows from excitation inductance 5e and leakage inductance 5d of transformer 5 through a parasitic diode 1a of first main MOS-FET 1, DC power source 3 and current resonance capacitor 4 to excitation inductance 5e to electrically charge current resonance capacitor 4.
When first main MOS-FET 1 is turned on at point t3 in time under the off-condition of second main MOS-FET 2, electric current ICi decreasingly flows from leakage inductance 5d of transformer 5 through first main MOS-FET 1, DC power source 3, current resonance capacitor 4 and excitation inductance 5e of transformer 5. Upon completion of energy release from leakage and excitation inductances 5d and 5e of transformer 5 at point t4 in time, electric current ICi flowing through current resonance capacitor 4 becomes substantially zero.
When electric current ICi through current resonance capacitor 4 has become nearly zero at point t4, charging current ICi for current resonance capacitor 4 starts flowing from DC power source 3 through first main MOS-FET 1, leakage and excitation inductances 5d and 5e of transformer 5, current resonance capacitor 4 to DC power source 3. In other words, during the period from point t4 to t5, electric current ICi flows through current resonance capacitor 4 in the adverse direction from that during the period from points t1 to t4 to reset magnetic flux produced in primary winding 6a of transformer 5.
When first main MOS-FET 1 is turned from on to off at point t5 under the off-condition of second main MOS-FET 2, electric current flows from current resonance capacitor 4 through voltage pseudo resonance capacitor 7, leakage and excitation inductances 5d and 5e of transformer 5 to current resonance capacitor 4 to discharge voltage pseudo resonance capacitor 7, while elevating voltage VQ1 between drain and source terminals of first main MOS-FET 1, and decreasing voltage VQ2 between drain and source terminals of second main MOS-FET 2.
When electric discharge from voltage pseudo resonance capacitor 7 is completed at point t6 under the off-condition of both first and second main MOS-FETs 1 and 2, voltage VQ2 between drain and source terminals of second main MOS-FET 2 comes to approximately zero, and at the same time, voltage VQ1 between drain and source terminals of first main MOS-FET 1 becomes equal to power voltage E. At this time, electric current ICi flows from excitation inductance 5e of transformer 5 through current resonance capacitor 4, parasitic diode 2a of second main MOS-FET 2 and leakage inductance 5d to excitation inductance 5e. 
When second main MOS-FET 2 is turned on at point t7 under the off-condition of first main MOS-FET 1, electric current flows from excitation inductance 5e of transformer 5 through current resonance capacitor 4, second main MOS-FET 2 to leakage inductance 5d of transformer 5.
At point t8, energy is transmitted from primary to secondary side of transformer 5, and a positive voltage appears on upper end of first secondary winding 5b of transformer 5. At this time, first output rectifying diode 8 of first rectifying smoother 10 is biased in the forward direction into the conductive condition, and therefore, voltage VD1 across first output rectifying diode 8 becomes approximately zero. Concurrently, due to resonance action by leakage and excitation inductances 5d and 5e and current resonance capacitor 4, circulation current branches from a path inclusive of excitation inductance 5e of transformer 5, current resonance capacitor 4, second main MOS-FET 2 and leakage inductance 5d of transformer 5 and flows through primary winding 6a of transformer 5. Also, due to resonance action by leakage inductance 5d of transformer 5 and current resonance capacitor 4, resonance current flows through primary winding 6a of transformer 5 along a path of primary winding 6a of transformer 5, current resonance capacitor 4, second main MOS-FET 2 and leakage inductance 5d of transformer 5. As a result, superimposed two circulation and resonance currents flow through primary winding 6a of transformer 5. Accordingly, sine wave-like load current ID1 starts flowing from first secondary winding 5b of transformer 5 through first output rectifying diode 8 while load current ID1 has the substantially same frequency as resonance frequency determined by leakage inductance 5d of transformer 5 and capacitance of current resonance capacitor 4.
When electric current ICi through current resonance capacitor 4 becomes nearly zero at point t9, circulation current flows from excitation and leakage inductances 5e and 5d of transformer 5 through second main MOS-FET 2 and current resonance capacitor 4, and at the same time, resonance current flows through primary winding 6a and current resonance capacitor 4 due to resonance action by leakage inductance 5d of transformer 5 and current resonance capacitor 4. Accordingly, superimposed circulation and resonance currents flow through current resonance capacitor 4 to discharge it. At this time, sine wave-like load current ID1 keeps flowing through first output rectifying diode 8 in secondary side until it comes to zero at point t10. During the period of time from point t8 to t10, voltage appearing on first secondary winding 5b of transformer 5 causes electric current to flow through first output rectifying diode 8 and first output smoothing capacitor 9 for commutation and smoothing to produce first DC output voltage VO1 at first DC output terminals 11 and 12.
At point t10 in time, circulation current flows from excitation inductance 5e of transformer 5 through leakage inductance 5d, second main MOS-FET 2 and current resonance capacitor 4 to accumulate energy in leakage and excitation inductances 5d and 5e of transformer 5. At the moment, voltage on first secondary winding 5b of transformer 5 is equal to or lower than first DC output voltage VO1 and inversely biasing voltage VD1 is applied on first output rectifying diode 8 which therefore is turned off into a non-conductive condition to stop electric current ID1 through first output rectifying diode 8. At time t11 after a cycle of first drive signal VG1 from main control circuit 15 has elapsed, second main MOS-FET 2 is turned from on to off while retaining first main MOS-FET 1 off, and from then on, the foregoing operations are repeated.
First output voltage detector 13 detects first DC output voltage VO1 on first DC output terminals 11 and 12 to produce error signal VE1, the difference between the detected voltage by detector 13 and reference voltage for prescribing first DC output voltage value VO1, and then error signal VE1 is transmitted to feedback signal input terminal FB of main control circuit 15 through photo-coupler 14 of light emitter 14a and light receiver 14b. Main control circuit 15 prepares first and second drive signals VG1 and VG2 of pulse frequency modulated based on voltage level of error signal VE1 forwarded from first output voltage detector 13 to feedback signal input terminal FB, and supplies first and second drive signals VG1 and VG2 to each gate terminal of first and second main MOS-FETs 1 and 2 to alternately turn them on and off with the frequency corresponding to voltage level of error signal VE1 from first output voltage detector 13, and thereby control toward and at a consistent value of first DC output voltage VO1 generated from first DC output terminals 11 and 12.
On-off operation of first and second main MOS-FETs 1 and 2 induces voltage on second secondary winding 5c of transformer 5 and also on second rectifying smoother 20. At this time, produced between both ends of second output smoothing capacitor 19 is DC voltage of the level accordant to turn ratio of first and second secondary windings 5b and 5c of transformer 5. DC voltage appearing between both ends of second output smoothing capacitor 19 is applied to step down chopper 27. Chopper controller 28 compares voltage VO2 between both ends of filter capacitor 26 with reference voltage for prescribing second output voltage value to produce a pulse-width modulated (PWM) signal VS2 based on an error signal, the difference between voltage VO2 and reference voltage. Step down chopper 27 controls the on-off operation of chopping MOS-FET 23 depending on PWM signal VS2 from chopper controller 28 to generate from second DC output terminals 21 and 22 second DC output voltage VO2 of a constant level lower than that of DC voltage applied to second output smoothing capacitor 19.
General switching power source devices of flyback or forward multi-output type are designed to control the DC output generated in secondary side by changing the on and off duty-ratio of main switching elements in primary side, and therefore, they are disadvantageous in changing period for transmitting electric power from primary to secondary side. Therefore, the above-mentioned duty ratio determined by DC voltage taken from one of secondary windings restricts electric power taken from the other of secondary windings so that output voltage from the other secondary winding is inconveniently reduced. On the contrary, resonant switching power source device of multi-output type can determine the period for supplying electric power from primary to secondary side of transformer 5 by resonance frequency given by current resonance capacitor 4 and leakage inductance 5d of transformer 5 in primary side, and therefore, almost no change arises in the period for supplying electric power from primary to secondary side of transformer 5 even though load connected to first DC output terminals 11 and 12 fluctuates. Consequently, whether load is big or small, the resonant switching power source device can produce electric power of necessary level from second secondary winding 5c of transformer 5 without drop in output voltage from second rectifying smoother 20. However, second rectifying smoother 20 may produce fluctuating output voltage because actually transformer 5 does not have an ideal electromagnetic coupling and/or due to impact on the output voltage by fluctuation in input voltage E or voltage drop in first rectifying smoother 10. For that reason, prior art resonant switching power source device shown in FIG. 6 employs step down chopper 27 to stabilize DC voltage from second rectifying smoother 20 and produce steady second DC output voltage VO2 from second DC output terminals 21 and 22. Specifically, step down chopper 27 at the subsequent stage of second rectifying smoother 20 can provide a resonant switching power source device of multi-output type capable of performing an ideal cross-regulation. The term “cross-regulation” means a fluctuation in output voltage produced under changing load or loads of other output in a prescribed range in a switching power source device of multi-output type.
Japanese Patent Disclosure No. 3-7062 demonstrates a resonant switching power source device which comprises a frequency modulator for modulating reference pulse signals in frequency into pulse array signals, a power transistor turned on and off by pulse array signals to control voltage applied on a primary winding of transformer, and a rectifying smoother provided in each of plural secondary windings for rectifying and smoothing an output from each secondary winding to output terminals. This power source device comprises a comparator as a primary control means for controlling frequency of pulse array signals produced from a frequency modulator in response to given output signals from rectifying smoother in secondary side. Also, a secondary control circuit is provided to turn a switching transistor on and off in response to given output signals from the secondary winding through a rectifying smoother in order to control the duty-cycle of pulse array voltages produced at the output side of switching transistor. This may thin an appropriate amount of pulse array voltages produced at the output side of switching transistor to adjust DC output voltage from secondary winding through rectifying smoother at a desired level.
Also, Japanese Patent Disclosure No. 2000-217356 shows a DC-DC converter of multi-output type which comprises a transformer having a primary winding and two secondary windings for power conversion, a field effect transistor connected to the primary winding of transformer of performing switching operation, a first voltage detector for detecting output voltage after stabilization of output from first secondary winding of transformer, a first pulse width modulator for comparing detection signal from first voltage detector with reference voltage to control pulse width of pulse control signals supplied to field effect transistor, a switch circuit connected to one end of second secondary winding of transformer, a second voltage detector for detecting rectified and smoothed output voltage from second secondary winding, a second pulse width modulator for comparing detected signals from second voltage detector with a reference voltage to modulate pulse width of pulse signals forwarded to switch circuit, and a synchronization circuit for synchronizing output from second pulse width modulator with output from first pulse width modulator. This DC-DC converter may control the on-time of switch circuit in secondary output line according to output voltage except the main feedback output to reduce power loss and stabilize output voltage even under large load fluctuation in the main feedback output.
Prior art resonant switching power source device shown in FIG. 6 may transmit energy to secondary side of transformer 5 by means of electric current ICi formed by merged and superimposed resonance current on circulation current flowing through primary winding 6a of transformer 5. In this case, as half-wave rectification is performed through first rectifying diode 8 in secondary side of transformer 5, resonance current joined into circulation current increases with increase in energy transmitted to secondary side. Since the only AC component of resonance current flows through primary winding 5a of transformer 5, resonance current of average value being approximately zero flows through primary winding 5a of transformer 5. In other words, positive and negative half cycles of this resonance current indicate substantially the same temporal area each other. Main control circuit 15 performs the on-off operation of first and second main MOS-FETs 1 and 2 while diverting electric current to parasitic diodes 1a and 2a of each main MOS-FET 1, 2 to control and alleviate fluctuation in voltage between drain and source terminals of main MOS-FETs 1 and 2 under the voltage pseudo condition. To transmit maximum energy to secondary side of transformer 5, while still retaining the voltage pseudo resonance condition, the time area of circulation current has to be substantially equal to the time area of resonance current flowing through primary winding 5a of transformer 5. Accordingly, to transmit more amount of energy to secondary side of transformer 5, more amount of circulation current has to flow through primary winding 6a of transformer 5. This results in increase in electric current flowing through primary winding 5a of transformer 5 and augmentation in self-heating by more amount of circulation current together with a large power conversion loss and a lowered power conversion efficiency.
To solve the foregoing problems, for example, a resonant switching power source device shown in FIG. 9, comprises an excitation reactor 34 connected in parallel to leakage and excitation inductances 5d and 5e of transformer 5, and excitation reactor 34 has an inductance value smaller than that of excitation inductance 5e of transformer 5 shown in FIG. 6. This circuitry allows a major amount of circulation current flowing in primary side of transformer 5 to divert through excitation reactor 34 to control effective value of electric current running through primary winding 5a of transformer 5. However, the power source device shown in FIG. 9 undesirably incorporates excitation reactor 34 which leads to increase in number of electric components and rise in cost for manufacture.
Also, the power source devices shown in the above-mentioned references comprise a switching element connected in any secondary line of transformer to adjust DC output voltage by the on-off operation of the switching element, and therefore, it has a drawback of current convergence in a specific output line upon turning-on of the switching element while disadvantageously providing a period of time of no current flow through other output lines than the specific output line. Especially, switching power source devices of boost type, which require higher output voltages, induce high voltages on secondary windings of transformer which result in rapid electric charge into output smoothing capacitors for very short charging time with large charging current. Inconveniently, this gives rise to a peak current appearing during the charging period of output smoothing capacitor, and peak current invites current concentration upon turning-on of secondary switching element with increase in power conversion loss and degradation in power conversion efficiency. Moreover, the foregoing current convergence causes uneven DC output voltages to develop from plural output terminals so that it makes difficult to separately generate steady DC output voltages of desired level from plural output terminals.
Therefore, an object of the present invention is to provide a resonant switching power source device which may reduce electric current flowing through a primary winding of a transformer. Another object of the present invention is to provide a resonant switching power source device which may control peak current flowing through each rectifying smoother in the secondary side for improvement in power conversion efficiency. Still another object of the present invention is to provide a resonant switching power source device which may independently generate stable DC output voltages of desired level from plural output terminals.